1. Field of the Invention
The present invention relates to a voltage generating circuit using a band gap type of constant voltage source formed in a Bi-CMOS semiconductor integrated circuit in which bipolar devices and complementary insulated gate devices are fabricated in the same substrate and, more particularly, to a voltage generating circuit for generating a reference potential for use with an emitter-coupled logic circuit (hereinafter abbreviated to an ECL circuit).
2. Description of the Related Art
FIG. 1 illustrates an example of an ECL circuit in which Q1 and Q2 designate a differential pair of NPN transistors having their bases respectively connected to receive a signal voltage Vin and a reference voltage VBB and their emitters connected together, Q3 a constant-current source NPN transistor having its collector connected to the emitters of transistors Q1 and Q2 and its base supplied with a reference voltage VCS, R1 and R2 resistors connected between V.sub.CC power supply and collectors of transistors Q1 and Q2, and R3 a resistor connected between the emitter of transistor Q3 and VEE power supply.
The above ECL circuit needs two types of reference potentials V.sub.BB and V.sub.CS, V.sub.BB being applied to the base of transistor Q2 as a threshold voltage which lies midway between a "1" logic level and a "0" logic level of ECL logic and Vcs being applied to the base of transistor Q3. The logical amplitude in ECL logic is low as about 0.8 volts and the allowable range of variability of reference potentials V.sub.BB and Vcs is small. Thus, a reference potential generating circuit is required which is small in the temperature dependence and power supply voltage dependence.
Heretofore, a band gap constant voltage circuit such as that as shown in FIG. 2 has been used as a voltage generating circuit for generating such reference potentials. As is well known, the constant voltage circuit uses such a Widlar circuit as shown in FIG. 3, in which Q1 to Q6 denote NPN transistors, R1 to R3 and R11 to R33 resistors, Vcc and V.sub.EE power supplies, Vcs and V.sub.BB reference potential outputs and A to C nodes.
Next, the principle of operation of the band gap constant voltage circuit and the Widlar circuit will be described with reference to FIGS. 4A, 4B and 5. In general, the base-to-emitter voltage V.sub.BE of a bipolar transistor has such temperature dependence as shown in FIG. 4A, the sign of which is negative. The thermal voltage VT of a semiconductor device is represented by k T/q (k=Boltzmann constant, T=absolute temperature and q=electronic charge) and has the temperature dependence the sign of which is positive as shown in FIG. 4B. In FIG. 5 which illustrates the principle of operation of the voltage generating circuit of FIG. 2, generation of k VT by VT generating circuit 91 and multiply-by-K circuit 92 and addition by adder circuit 94 of V.sub.BE from V.sub.BE generating circuit 93 and K VT will meet the following temperature compensation condition: EQU (dV.sub.BE /dT)+K dVT/dT)=0 (1)
The output potential Vout will be a constant potential with no temperature dependence which is given by EQU Vout=V.sub.BE +K VT (2)
In the Widlar circuit of FIG. 3, assuming that currents flowing through transistors Q1, Q2 and Q3 are I1, I2 and I3, respectively, diode saturation currents of transistors Q1 and Q2 are Is1 and Is2, respectively, and base currents of transistors are small enough to be neglected, then a voltage V1 across resistor R1 will be given by EQU V1=VT nI1 / Is1 EQU V1=I2R3+(VT n I2 / Is2)
A voltage V2 across resistor R2 will be given by ##EQU1##
Adder circuit 94 for adding V.sub.BE and K VT shown in FIG. 5 can be implemented by connecting to the base of transistor Q3 a low-potential end of resistor R2 across which voltage V2 is developed. A potential difference between the high-potential end of resistor R2 and the emitter of transistor Q3 is given by expression (2). The condition of expression (1) can be met by adjusting the emitter area ratio (Is1/Is2) of transistors Q1 and Q2, current ratio (I1/I2) and resistance ratio (R2/R3) in expression (3).
In the band gap constant voltage circuit shown in FIG. 2, resistor R33 serves as a bias resistor for transistors Q4 and Q5 as well as a current source of current I3. Also, transistors Q4 and Q5 serve as current sources of currents I1 and I2. Potential difference Vcs between node B and V.sub.EE potential point has no temperature dependence. If resistors R22 and R2 have the same resistance, then the same voltage as voltage V2 across resistor R2 will be developed across resistor R22. If currents I1 and I3 flowing through transistors Q6 and Q3 are adjusted to keep the same emitter current density, then the same base-to-emitter voltages V.sub.BE will be developed, which have the same temperature dependence. Thus, the potential difference V.sub.BB between V.sub.CC potential point and node A will have no temperature dependence as with the base-to-emitter voltage of transistor Q3.
However, a voltage across the resistance R3 varies to a greater extent than a power supply voltage so that the dependency of the current I3 upon the power supply voltage is greater. The base-to-emitter voltage V.sub.BE of the transistor Q3 increases with an increasing current and an output potential Vout also has a power supply voltage dependency.
Thus, as the current I3 reveals such a power supply voltage dependency, so the base-to-emitter voltage V.sub.BE of the transistor Q3 also reveals the power supply voltage dependency. As appreciated from the expression (3), the output voltage Vout has the power supply voltage dependency.
Further, the temperature coefficient d V.sub.BE /dT of the base-to-emitter voltage V.sub.BE of the bipolar transistor varies due to a collector current (when the collector current increases, the absolute value of the temperature coefficient V.sub.BE /dT decreases). For this reason, the temperature requirement as represented by the expression (1) varies due to a variation of electric current I.sub.3 caused by a variation of the power supply voltage. Thus the output voltage Vout is not temperature-compensated over a broader power source voltage range and has a temperature dependency.
That is, problems arise with the prior art band gap constant voltage circuit shown in FIG. 2 in that, as shown in FIGS. 6A and 6B, current I3 increases with increasing power supply voltage (voltage between V.sub.CC potential and V.sub.EE potential), currents I1 and I2 increase with an increase in the potential at node C, the temperature compensation condition represented by expression (1) becomes unsatisfied, and voltage V.sub.BB between node A and Vcc potential and voltage Vcs between node B and V.sub.EE potential increase.
To eliminate the above problems, such a band-gap type voltage regulator circuit as shown in FIG. 7 has been used. In FIG. 7, a resistor Rc is connected between the collector of transistor Q3 and resistor R33 and a PNP transistor Qc has its collector connected to V.sub.EE and its base emitter path connected across resistor Rc so as to clamp the voltage across resistor Rc to hold current I3 constant. According to such a band-gap voltage regulator, the temperature compensation condition is satisfied over a wide range of the supply voltage so that output voltage Vout will have no temperature dependence.
A problem arises, however, in the case where PNP transistor Qc is fabricated in a bipolar integrated circuit along with NPN transistors Q1 to Q6 in that additional manufacturing steps are required. This will increase manufacturing cost and decrease yield.
As described above, the problems with the prior art voltage regulator are an increase in manufacturing steps, an increase in cost and a decrease in yield which result from the use of a PNP transistor for satisfying the temperature compensation condition over a wide range of the power supply voltage to produce an output voltage with no temperature dependence.